The switching frequency of a main switching device in a power converter can be a key parameter which affects the electrical performance as well as the cost of the power converter. The size of passive elements, especially the magnetic elements, can be reduced by increasing the switching frequency of the converters. An input choke of a boost converter or an isolation transformer of a DC-DC converter can be used as examples of such magnetic elements. This reduction can have a direct and significant impact on the overall cost of the converter.
Known power converter applications having cost-efficiency as the main priority may greatly benefit from a possibility to increase the switching frequency. For example, in power converters for data centres or base stations of telecommunication applications, the switching frequency can be in the range of 200 kHz to 600 kHz. Such a range can offer a large potential for increasing the switching frequency, and, thus, also for reducing the size of the magnetic components. Reduction in the size of the magnetic component can, as mentioned, lead to significant cost savings and an increase in power density. Further, by increasing the switching frequency to a certain range, such as 20 kHz or above, low-cost and low-core-loss materials, such as soft ferrite, can be used for the magnetic elements of the power converters.
However, a large increase in the switching frequency of an existing power converter can include a trade-off. For example, an increase from a few kHz to tens of kHz can result in very high switching losses.
FIG. 1a shows an exemplary switching waveform of a known power converter, for example a 750-W boost converter with a 400-V output. In FIG. 1a, the voltage VCds,S and the current iS of the switching device are shown. High current stresses 10 in the current iS are induced at turn-on at point 11 by reverse recovery current of the main diode and the high switching speed of the main switching device. FIG. 1a also shows high voltage stress 12 at turn-off at point 13, induced by high switching speed and the parasitic inductance of the circuit.
FIG. 1b shows corresponding switching trajectories of the known power boosted illustrated in FIG. 1a. The turn-on trajectory 14 and the turn-off trajectory 15, together with the current axis and the voltage axis, enclose areas which can correspond with switching losses in terms of energy dissipated in turn-on and turn-off actions of the main switching device. Thus, the switching losses increase linearly with the switching frequency.
Higher switching losses generate more heat, and a more powerful cooling system or a larger heat sink may be specified for extracting the heat efficiently and keeping the semiconductors from overheating. As a result, the power density and the power efficiency of the converter may degrade, and the cost savings gained in the magnetic parts may be nullified by the increased cooling system costs.
FIGS. 2a-2c illustrate quasi-resonant switches in accordance with known implementations. So called soft-switching can be used to achieve more better results, e.g., high switching frequency and low switching loss simultaneously. In order to change the switching of a converter from known hard-switching to soft-switching, at least two approaches may be used.
Soft-switching can be achieved by using a quasi-resonant switch, e.g., by replacing a known PWM switching cell, such as the one illustrated in FIG. 2a, by a quasi-resonant switching cell, such as the one illustrated in FIG. 2b or 2c. FIG. 2b shows a half-wave zero-current resonant switching cell whereas FIG. 2c shows a full-wave zero-current resonant switching cell.
A quasi-resonant switch can switch under zero-current turn-on and zero-voltage turn-off conditions. However, an additional resonant component and diode are connected in series with the main switch, which can increase the conducting state losses. Moreover, the main switch may suffer from either over-voltage or over-current stress. The stresses can be increased with the power rating of the converter. Compared with a hard-switching converter, a semiconductor switch with a higher rating may be specified. Higher rating, in turn, may increase the cost of the switch.
Another exemplary approach is to use an auxiliary circuit, a snubber, to assist the main switch to perform either zero-voltage or zero-current switching. A snubber can be defined as a circuit that is able to modify turn-on and/or turn-off switching trajectories of semiconductor switches and to reduce, or even eliminate, switching losses by processing a small amount of reactive power. FIG. 3 shows an exemplary block diagram of a snubber in a power converter according to a known implementation.
The rates of change di/dt and dv/dt in switching events can be lowered by resonant actions of the snubber. Oscillations induced by the switching actions and parasitic capacitors and inductors can also be reduced. As a result, EMI problems can be reduced.
Different snubber circuits have already been published in various scientific papers and patent publications. The proposals can be differentiated from each other mainly by achieving zero-voltage or zero-current switching and by the reset circuit of the snubber. U.S. Pat. No. 6,987,675B2, U.S. Pat. No. 6,028,418A, U.S. Pat. No. 5,313,382A, U.S. Pat. No. 6,236,191B1, U.S. Pat. No. 5,959,438A, U.S. Pat. No. 5,418,704A, US Patent Application US20020047693A1, and South Korean Patent Application KR20040054088A disclose some exemplary approaches for implementing snubber circuits.